Impact of Physical Layer Impairments on C+L-band Metro Networks

Filipa F. Gomes

1

, Lu

´

ıs G. Cancela

1,2

and Jo

˜

ao L. Rebola

1,2

1

Optical Communications and Photonics Group, Instituto de Telecomunicac¸

˜

oes, Lisbon, Portugal

2

Department of Information Science and Technology, Instituto Universit

´

ario de Lisboa (ISCTE-IUL), Portugal

Keywords:

ASE Noise, In-band Crosstalk, Metro Networks, Multi-band Transmission, Nonlinear Interference Noise.

Abstract:

To ensure future data trafﬁc capacities, multi-band transmission in optical networks is being proposed, by

adding wavelength division multiplexing channels in the L-band to the existing C-band channels. In this

work, we study the ﬁlterless drop and waste ampliﬁed and unampliﬁed solutions used in metro networks with

L-band tributary nodes. The cost and power consumption of these nodes is evaluated, as well as the impact of

several physical layer impairments on the network performance. In ampliﬁed solutions, we show that L-band

networks with 9 tributary nodes are feasible, being the optical signal-to-noise ratio achieved when considering

ampliﬁers noise and ﬁber non-linear interference above 14 dB. In unampliﬁed solutions, to fullﬁl the power

budget, frequency reuse should be used between spans. We show that the in-band crosstalk caused by this

frequency reuse leads to 1.5 dB power budget degradation, for a crosstalk level of -20 dB.

1 INTRODUCTION

To offer the best Internet experience, nowadays, it is

necessary to resort to cloud services which, in the

future, will probably saturate the transport capacity

of existing metro networks. Several solutions based

on multi-band transmission are being investigated to

overcome this possible capacity crunch (Ferrari et al.,

2020), but the introduction of the L-band (by adding

wavelength division multiplexing (WDM) channels)

opens the opportunity for cheaper solutions than plac-

ing several ﬁbres in parallel.

Filterless solutions, both ampliﬁed and unampli-

ﬁed, are being proposed to reduce even more the cost

of upgrading the optical networks to the L-band (Em-

merich et al., 2020b), (Paolucci et al., 2020). In

particular, the ﬁlterless solution is usually applied to

networks with horseshoe topology (Emmerich et al.,

2020b). In (Paolucci et al., 2020), the L-band ﬁlter-

less unampliﬁed solution is proposed, but due to fre-

quency reuse, in-band crosstalk becomes a relevant

impairment that degrades the network performance.

As this solution is only suitable for short-reach spans,

noise from the ampliﬁers and ﬁber non-linearities can

be neglected. Due to the impossibility of reusing fre-

quencies in the ampliﬁed solution, ampliﬁed sponta-

neous emission (ASE) noise and ﬁber non-linear in-

terference (NLI) become the main impairments that

degrade the network performance.

In all these works, regarding ﬁlterless L-band

transmission, we have seen that they resort on ex-

perimental results and simulations to study the up-

grade of the metro area networks. However, an an-

alytical study of the ﬁlterless solutions has not been

performed in these works.

In this work, we analyse the node components and

architectures for the C+L-band, and their cost and

power consumption are estimated. A simple analyt-

ical formalism of the various physical layer impair-

ments (PLIs) that affect the performance of the ﬁl-

terless solution, either ampliﬁed or unampliﬁed, in

the horseshoe topology is provided. The impact of

ASE noise and NLI on the optical signal-to-noise ra-

tio (OSNR) of the horsehoe topology is studied for

ﬁlterless L-band ampliﬁed solutions and the inﬂuence

of in-band crosstalk on the electrical signal-to-noise

ratio (SNR) is assessed for unampliﬁed solutions.

This work is organized as follows. In section 2,

we review the L-band node architectures ampliﬁed

and unampliﬁed solutions for the C+L transmission.

In section 3, we perform an analysis regarding the

L-band hardware cost and power consumption. In

section 4, the several PLIs that impair the L-band

nodes network performance are studied for the ampli-

ﬁed and unampliﬁed solutions. In section 5, the ﬁnal

conclusions are drawn.

134

Gomes, F., Cancela, L. and Rebola, J.

Impact of Physical Layer Impairments on C+L-band Metro Networks.

DOI: 10.5220/0010909900003121

In Proceedings of the 10th International Conference on Photonics, Optics and Laser Technology (PHOTOPTICS 2022), pages 134-143

ISBN: 978-989-758-554-8; ISSN: 2184-4364

Copyright

c

2022 by SCITEPRESS – Science and Technology Publications, Lda. All rights reserved

2 NODE ARCHITECTURE FOR

C+L-BAND

Several solutions regarding metro networks with C+L

band transmission rely on the horseshoe network

topology to manage the access/metro network data

trafﬁc (Paolucci et al., 2020). The horseshoe topol-

ogy is usually composed of two hub nodes (which are

nodes that communicate in a meshed topology with

other hub nodes) and several tributary nodes (which

are nodes that connect with the hub nodes or other

tributary nodes, and perform the interface with ac-

cess metro networks) (Paolucci et al., 2020), (Em-

merich et al., 2020a). As shown in Fig. 1, a C+L

tributary node architecture has two independent struc-

tures, one for C-band signals and other for L-band sig-

nals (Emmerich et al., 2020a). These two bands are

separated/aggregated by a C/L band ﬁlter at the in-

put/output of the node, which typically provides high

bands isolation and low insertion losses, introducing a

300 to 500 GHz gap between the C and L bands. The

separated L-band solution causes a minimum disrup-

tion with pre-existing metro networks working only

in the C-band (Emmerich et al., 2020a).

The node conﬁguration used for the C-band,

which is shown at the top of Fig. 1 (blue color),

can be implemented with various nodes architectures

such as ﬁlterless drop and waste (FD&W), ﬁxed op-

tical add/drop multiplexers (FOADM) and reconﬁg-

urable optical add/drop multiplexer (ROADM) solu-

tions (Emmerich et al., 2020a).

Figure 1: Tributary C+L-band node conﬁguration.

The node conﬁguration for the L-band is shown at

the bottom of Fig. 1 (orange color). It is based only on

the FD&W solution, whose architecture relies only on

splitter/couplers (S/Cs), and can be either ampliﬁed or

unampliﬁed (Emmerich et al., 2020a). This solution

has no ﬁltering in the tributary nodes, which means

that when a wavelength is added to the network, it re-

mains in the network until the hub node removes it

(ampliﬁed solution) or until it is reused once again in

the network (unampliﬁed solution) (Emmerich et al.,

2020b), (Paolucci et al., 2020). The hub node L-band

structure typically uses ROADMs as in the C-band

structure counterpart (Hernandez et al., 2020).

2.1 L-band Ampliﬁed Solution

In the ampliﬁed solution, there are optical ampliﬁers

(OAs) inside the node before and after the FD&W

block, see Fig. 1 (Paolucci et al., 2020).

Fig. 2 presents a possible wavelength planning

for the ampliﬁed solution in the L-band in a horse-

shoe topology, where each tributary node has a spe-

ciﬁc wavelength to communicate with the hub node.

This solution does not allow the reuse of frequencies

inside the network since each wavelength can only

be extracted by the hub node (represented in pink in

Fig. 2). The tributary nodes (represented in green in

Fig. 2) do not have such capacity since they are based

on S/Cs. In Fig. 2, the pink wavelength λ

5

is as-

signed to the connection between the hub node 1 and

the tributary node E, but node E can not remove this

wavelength, which is only removed from the network

at the hub node 2. This path is represented by the

dashed pink line, and represents a wasted wavelength

for nodes F to I (Emmerich et al., 2020a).

Figure 2: Ampliﬁed ﬁlterless solution with no frequency

reuse for the horseshoe topology. Only the L-band wave-

length assignment for the downlink transmission is de-

picted.

2.2 L-band Unampliﬁed Solution

The main goal of the L-band unampliﬁed solution is

to reduce the nodes cost by not using OAs. This so-

lution limits the L-band transmission distance to ad-

jacent nodes or near adjacent nodes (Emmerich et al.,

2020a).

Fig. 3 shows an example of a wavelengh assign-

ment for the unampliﬁed solution, to illustrate a wave-

length reuse possibility. As shown in Fig. 3, the wave-

length λ

2

is reused in the next-hop and λ

1

is reused

after one hop. As the WDM channel is not ampliﬁed,

Impact of Physical Layer Impairments on C+L-band Metro Networks

135

it arrives to the next node with very low power, mean-

ing that a new channel with the same wavelength can

be used. However, the (even weak) power leakage on

node H can impair the new channel leading to in-band

crosstalk. (Emmerich et al., 2020b).

Figure 3: Unampliﬁed ﬁlterless solution with frequency

reuse for the horseshoe topology. Only the L-band wave-

length assignment for the downlink transmission is de-

picted.

2.3 L-band Nodes Architectures

In the L-band, the tributary nodes use a FD&W ar-

chitecture, while the hub nodes have usually a color-

less and directionless (CD) ROADM route and select

(R&S) architecture, as shown, respectively, in Figs. 4

and 5.

In Fig. 4, the tributary node architecture consid-

ers the transmission of 70 WDM channels with a 100

GHz channel spacing in each ﬁber direction. For two

directions and a add/drop (A/D) ratio of 20%, the

number of required coherent detection transponders

in each L-band node is 28. Thus, beside considering

two C/L ﬁlters to separate/aggregate the two bands

and two ampliﬁers per node (in the case of the am-

pliﬁed solution), each tributary FD&W node has four

S/Cs 1:2, four S/C 1:16, each with 4 unused channels

and 28 transponders, as shown in Fig. 4.

Figure 4: 2-Degree FD&W tributary node.

In Fig. 5, we consider that each 5-degree hub node

is connected to 3 different hub nodes and two tribu-

tary nodes. So, for an A/D ratio of 20%, a total of

112 wavelengths must be added/dropped in each hub

node. As wavelength selective switches (WSSs) with

maximum dimension of 1:20 are considered (Hernan-

dez et al., 2020), it is necessary to use 6 cards in

the A/D structure to achieve this number of wave-

lengths. To connect each A/D card to the ROADM

input/output directions, express WSSs 1:20 are con-

sidered. So, in Fig. 5, the hub node has ﬁve WSSs

1:20 at the input and output stages, twelve WSSs 1:20

in the A/D structure, six WSSs 1:8 at the drop section,

six S/Cs 1:8 at the add section and 112 transponders.

Figure 5: 5-Degree CD ROADM R&S hub node.

3 HARDWARE COST AND

POWER CONSUMPTION

In this section, an analysis regarding the cost, power

consumption and total offered capacity of an horse-

shoe topology with 2 hub nodes and 9 tributary nodes

(as shown in Fig. 2) is performed and presented in

Tab. 1. The analysis is performed considering that

the cost and power consumption of the hardware com-

ponents in the L-band is twice the C-band compo-

nents cost, which is taken from (Hernandez et al.,

2020). The cost is normalized to the cost of a 10 Gbps

transponder, whose price is around 330 e (Hernandez

et al., 2020).

For comparison purposes, the cost and power con-

sumption of the same horseshoe network have been

calculated for the C-band FD&W solution consider-

ing the components presented in (Hernandez et al.,

2020), and the same A/D ratio of 20%.

The total capacity per ﬁber reaching the tributary

and hub nodes in the L-band is, respectively, 7 Tbps

and 28 Tbps, determined from the number of WDM

channels and the transponder capacity, which is, re-

spectively, 70 channels and 100 Gbps for the tributary

PHOTOPTICS 2022 - 10th International Conference on Photonics, Optics and Laser Technology

136

node and, respectively, 140 channels and 200 Gbps

between hub nodes. The total capacity per node pre-

sented corresponds to the capacity per A/D structure.

So, the capacity per tributary node is 2.8 Tbps and the

average total capacity per hub node is calculated con-

sidering that 2/5 of the transponders are working at

100 Gbps (to communicate with tributary nodes) and

the other 3/5 are at 200 Gbps (to communicate with

other hub nodes). Hence, the average total capacity

per hub node is 17.9 Tbps.

Table 1: Total cost, power consumption and capacity of the

hub and tributary nodes and per horseshoe network.

Cost/Consumption Node architecture

Hub

Normalized cost/node 1952.44

Power consumption/node [kW] 18.96

Total capacity/node [Tbps] 17.9

Trib. Ampliﬁed

Normalized cost/node 282.2

Power consumption/node [kW] 3.43

Total capacity/node [Tbps] 2.8

Total cost/horseshoe [k] 6.45

Total pow.cons./horseshoe [kW] 68.75

Trib. Unampliﬁed

Normalized cost/node 280.95

Power consumption/node [kW] 3.36

Total capacity/node [Tbps] 2.8

Total cost/horseshoe [k] 6.43

Total pow.cons./horseshoe [kW] 68.15

The tributary node unampliﬁed solution is cheaper

(less 1.25 of normalized cost) and spends less power

per node (around 20 W) than the tributary node ampli-

ﬁed solution. The difference between these solutions

is only due to the existence or not of OAs. The com-

ponents that most contribute, around 99%, to the ﬁnal

cost of the node are the transponders. The normalized

cost per tributary node is around 3.5 times higher than

in the C-band. This occurs because, in the L-band,

the number of transponders is almost the double. The

power consumption of the tributary node is 3.4 kW,

which is around 1.7 times higher than the consump-

tion of the FD&W for the C-band. Other components,

such as the S/Cs have a small contribution to the ﬁnal

cost.

The hub node has a normalized cost around 3.5

times higher than the C-band node cost. The power

consumption is also around 3.2 times higher than the

corresponding C-band node power consumption.

So, the estimated total network cost using FD&W

with ampliﬁed and unampliﬁed soutions is 2134 ke

and 2128.3 ke, respectively, when the components

are twice expensive, which means that the unampli-

ﬁed solution is 0.3% cheaper than the ampliﬁed one.

4 PHYSICAL LAYER

IMPAIRMENTS IN THE

L-BAND

In this section, the analysis of several PLIs, origi-

nated inside the nodes as well as in ﬁber transmission,

such as insertion losses, ASE noise, NLI noise and in-

band crosstalk is performed, considering quadrature

phase-shift keying (QPSK) signal transmission in the

L-band. The PLIs impact is quantiﬁed by the optical

signal-to-noise-ratio (OSNR) given by (Semrau et al.,

2019):

osnr =

p

Rx

p

n,ASE

+ p

NLI

(1)

where p

Rx

is the average signal power at the optical

receiver input, p

n,ASE

is the accumulated ASE noise

power and p

NLI

is the NLI power at the receiver in-

put. The following parameters are considered: the

ampliﬁers noise ﬁgure (F

n

) is 6.5 dB, at most 0.5 dB

higher than in C-band (Correia et al., 2021). The

ﬁber non-linear coefﬁcient in the L-band, γ = 1.2

W/km, is slightly lower than in the C-band (Semrau

et al., 2018), and the attenuation coefﬁcient is α=0.25

dB/km, slightly higher than in the C-band. The chan-

nel bandwidth of 26.75 GHz is equal to the symbol

rate, R

s

, and the WDM channel spacing is 100 GHz.

Fig. 6 represents the spectral occupancy of the

70 channels belonging to the L-band and of the

40 C-band channels, considering the International

Telecommunication Union (ITU)-T 100 GHz channel

grid (Telecommunication Standardization Sector of

ITU, 2020). The L-band central frequency is 188

THz, while, in the C-band, is 193.7 THz. A band gap

of 300 GHz between the C and L bands is considered,

which contains 3 unused channels.

Figure 6: WDM signal frequencies for the C+L-band with

100 GHz channel spacing following the ITU-T grid, consid-

ering 40 channels for C-band and 70 channels for L-band.

Regarding the optical ﬁltering effect in the L-

band, only the WSSs in the hub nodes contribute to

this effect. Hence, the effect of the optical ﬁltering is

neglected in the following analysis.

Impact of Physical Layer Impairments on C+L-band Metro Networks

137

4.1 Insertion Losses in Ampliﬁed and

Unampliﬁed Solutions

In this section, we present the insertion losses of the

ampliﬁed and unampliﬁed solutions. The losses of

each hardware element are those considered in (Her-

nandez et al., 2020). The WSS losses, which are in-

dependent of the WSS size are 7 dB (Sequeira et al.,

2021). The insertion losses considered for the S/Cs

1:2, 1:4, 1:8 and 1:16, are respectively, 4, 7, 11 and

15 dB (Inc., 2019). The losses of the C/L band ﬁl-

ters are around 0.6 dB (Fiberdyne labs, 2021). The

connectors and splices losses are neglected.

In Tab. 2, the insertion losses inside the add, drop

and express paths of the tributary and hub nodes are

shown. The ﬁber losses after 10 km are 2.5 dB. The

hub node leads to the highest losses, 25 dB. For the

tributary node, the add and drop paths losses are the

same, 19.6 dB.

Table 2: Insertion losses in the L-band, of the tributary node

the hub node.

Tributary node

Add

path

S/C 16:1+S/C 2:1+C/L ﬁlter=19.6 dB

Drop

path

S/C 1:2+S/C 1:16+C/L ﬁlter=19.6 dB

Express

path

S/C 1:2 +S/C 2:1 + 2 ×C/L ﬁlter=9.2 dB

Hub node

Add

path

WSS 20:1+S/C 1:8+WSS 20:1=25 dB

Drop

path

WSS 1:20 +WSS 1:8+WSS 1:20=21 dB

Express

path

WSS 1:20 +WSS 20:1=14 dB

10 km ﬁber 2.5 dB

The ampliﬁer gains are designed to perfectly com-

pensate the insertion losses of the worst path inside

the node, the add path. For tributary nodes, only pre-

ampliﬁcation is used and the ampliﬁer gain is 22.1 dB

to compensate the losses of the 10 km span and node

losses.

4.2 ASE Noise in the Ampliﬁed Solution

In this section, the accumulated ASE noise power is

calculated for the ampliﬁed solution in the L-band,

considering two worst paths in the horseshoe network,

1) channel insertion at the hub node 1 and drop at

node I in Fig. 2, and 2) channel insertion at node

A and drop at hub node 2. The accumulation of ASE

noise is quantiﬁed by the OSNR using (1), ﬁrstly, con-

sidering the absence of NLI.

The accumulated ASE noise power is given by:

p

n,ASE

= 2 ×

n

spans

∑

i=1

[p

ASE, pre,i

+ p

ASE, pos,i

] (2)

where n

spans

is the number of spans, p

ASE, pre,i

is the

ASE noise power per polarization generated at each i-

th pre-ampliﬁer output and p

ASE, pos,i

is the ASE noise

power per polarization generated at each i-th post-

ampliﬁer output. The ASE noise power of each am-

pliﬁer is given by

p

ASE,m,i

=

f

n,m,i

2

× (g

m,i

− 1) × h ×ν

0

× R

s

(3)

where m=pre or pos, h is the Planck’s constant and

ν

0

is the optical frequency. The ASE noise power as a

function of the channel frequency (in lowpass equiv-

alent deﬁnition) after nine spans of the horseshoe net-

work for the cases of channel insertion at tributary

node A and hub node 1 is shown in Fig. 7. The ASE

noise power increases slightly with frequency and the

difference between ﬁrst and last channel frequencies

is less than 0.3 dB. The ASE noise power is about 0.7

dB higher, when the channel insertion is in the trib-

utary node in comparison with insertion in the hub

node.

Figure 7: Accumulated ASE noise power as a function of

the channel frequency after 9 spans of the horseshoe net-

work for channel insertion in hub 1 and tributary A nodes.

Fig. 8 shows the OSNR evolution along the net-

work, for both cases of signal insertion, for the chan-

nel frequency of 188 THz. When signal insertion oc-

curs at node 1, the OSNR is signiﬁcantly degraded

as the signal crosses the nodes, decreasing 9.3 dB.

For signal insertion at node A, the OSNR decreases

7.1 dB. The minimum reference OSNR is 8.5 dB to

achieve a target BER 4 × 10

−3

(Sambo et al., 2020).

Fig. 8 shows that, for both cases of signal insertion,

the OSNR achieved at the optical receiver input pro-

vides a good safety margin in relation to the minimum

OSNR.

4.3 NLI in the Ampliﬁed Solution

In this section, the NLI noise power and the associated

OSNRs are studied for the ampliﬁed solution. To per-

PHOTOPTICS 2022 - 10th International Conference on Photonics, Optics and Laser Technology

138

Figure 8: OSNR in the horseshoe topology, for signal inser-

tion at the tributary node A and signal insertion at the hub

node 1 for p

Rx

=0 dBm.

form these studies, the NLI calculation and the OSNR

formulation must take into account the presence of

inter-channel Stimulated Raman Scattering (SRS). In

this section, the Gaussian noise model presented in

(Semrau et al., 2019) is considered, where the effect

of the SRS in the NLI noise is taken into account in

simpliﬁed closed-form expressions.

The NLI parameter in the L-band, used in the cal-

culation of the OSNR, is given by (Semrau et al.,

2019)

η( f

i

) ≈

n

spans

∑

j=1

P

i, j

P

i

2

[η

SPM, j

( f

i

)n

ε

spans

+ η

XPM, j

( f

i

)]

(4)

where η

SPM, j

( f

i

) is the self-phase modulation (SPM)

contribution generated in j-th span at frequency f

i

,

η

XPM, j

( f

i

) is the total cross-phase modulation (XPM)

contribution generated in j-th span, and P

i, j

is the

power of i-th channel launched into the j-th span, and

ε is the coherent factor of NLI accumulation along

multiple ﬁber spans. In this work, as we consider

exact loss compensation, P

i, j

=P

i

, and NLI coherent

accumulation is assumed. The SPM contribution is

given by (Semrau et al., 2019):

(5)

η

SPM

( f

i

) ≈

4

9

γ

2

B

2

i

π

Φ

i

α(2α + α)

·

T

i

− α

2

α

arcsinh

Φ

i

B

2

i

πα

+

A

2

− T

i

A

arcsinh

Φ

i

B

2

i

πA

where Φ

i

=

3

2

π

2

(β

2

+ 2π β

3

f

i

), β

2

is the group-

velocity dispersion and β

3

is the higher order dis-

persion parameter (Agrawal, 2010), A= α + α and

T

i

= (α+α−P

tot

C

r

f

i

)

2

, where P

tot

is the WDM signal

launch power, B

i

is the bandwidth of the i-th channel

and C

r

is the slope of the linear regression of the nor-

malized Raman gain spectrum (Semrau et al., 2019).

The total XPM contribution is given by (Semrau et al.,

2019)

η

XPM

( f

i

) ≈

32

27

N

ch

∑

k=1,k6=i

P

k

P

i

2

γ

2

B

k

Φ

i,k

α(2α + α)

·

T

k

− α

2

α

arctan

Φ

i,k

B

i

α

+

A

2

− T

k

A

arctan

Φ

i,k

B

i

A

(6)

where Φ

i,k

= 2π

2

( f

k

− f

i

)[β

2

+ πβ

3

( f

i

+ f

k

)]. In this

work, we consider the simpliﬁed case of α = α.

For the NLI study, C

r

=0.028 1/W/km/THz, the

dispersion parameter is 17 ps/nm/km and the disper-

sion slope is ps/nm

2

/km. In Fig. 9 (a), the NLI param-

eter as a function of the WDM channels frequency is

shown, for several input powers per channel, consid-

ering the horseshoe topology with 9 tributary nodes

and 10 km spans. Fig. 9 (a) reveals that, for higher

channel input power, the higher the NLI parameter

and the steepest its tilt, due to the stronger power

transfer from higher to lower frequencies caused by

SRS.

In Fig. 9 (b), the corresponding NLI power as a

function of the WDM channels frequency is shown.

The NLI power is determined by p

NLI

=

P

3

rx

η

(Semrau

et al., 2019), As can be observed in Fig. 9 (b), the

NLI power increases almost 5 dB, for a 4 dBm power

channel, as the channel frequency decreases from the

highest to the lowest channels. Also, it can be ob-

served that, as the input power decreases, the NLI

power increase is smoother from the highest to the

lowest frequency channel. For the -2 dBm channel

power, the NLI power remains practically constant for

all frequency channels due to the reduced SRS. Notice

that the results presented for the NLI power are inde-

pendent on where the signal is inserted in the network.

The OSNR as a function of the channel frequency

after 9 spans considering the ASE noise power and

NLI is shown in Fig. 10, for several input chan-

nel powers, including the optimal power per chan-

nel, which leads to the best OSNR, and the maximum

power per channel to comply the minimum OSNR,

for the worst channel. Fig. 10 corresponds to the case

of: (a) channel insertion at the tributary node A and

(b) channel insertion at the hub node 1. The OSNR is

calculated using Eq. (1).

As can be seen in Fig. 10, for the input power

per channel of -2 and 0 dBm, the NLI power tilt

has practically no inﬂuence, so an almost constant

Impact of Physical Layer Impairments on C+L-band Metro Networks

139

Figure 9: NLI parameter and p

NLI

as a function of the chan-

nels frequency after nine 10 km spans considering coherent

NLI accumulation, for several input powers per channel.

OSNR along all channels is observed. For input pow-

ers higher than 0 dBm, we observe that the OSNR

increases as the channel frequency increases due to

the higher NLI power contribution to the OSNR slope

variation. So, we can state that there is an input chan-

nel power that gives an optimum OSNR. This opti-

mum OSNR is obtained by evaluating the best possi-

ble OSNR, with the highest ﬂatness along the WDM

signal bandwidth (slope as close to zero as possible).

Hence, the optimal input power that leads to the best

OSNR is -1 and -0.8 dBm, respectively, in Fig. 10

(a) and (b), corresponding to the optimal OSNRs of

14.5 and 14 dB. It can be observed that the optimal

OSNR is practically constant along frequency ensur-

ing the same received quality for all WDM channels.

When the signal is inserted in the hub node, the op-

timum OSNRs are around 0.5 dB lower than in the

case of channel insertion in the tributary node due to

the higher ASE noise power added by the hub node.

Finally, it can be observed that all channel powers

considered in Fig. 10 comply with the 8.5 dB mini-

Figure 10: OSNR as a function of the channel frequency

after 9 spans considering the ASE noise power and NLI, for

several input channel powers and insertion channel in the

(a) tributary node A and (b) hub node 1.

mum OSNR required, except for the input power of

4 dBm, where this value is not achieved in the low-

est channel frequencies. The maximum input power

that ensures the minimum OSNR for all WDM chan-

nels is also represented in Fig. 10 and corresponds to

3.8 dBm, for both cases of channel insertion. There-

fore, the difference between the optimal OSNR and

the minimum OSNR is at least 5.5 dB higher, which

ensures enough margin to include other PLIs. For ex-

ample, if we consider a minimum system margin of

3 dB and an extra power penalty due to other trans-

mission effects of 1.5 dB, like components aging, the

minimum OSNR is still accomplished.

4.4 In-band Crosstalk in the

Unampliﬁed Solution

In this subsection, the analysis of PLIs, considering

the signal transmission in the L-band in a horseshoe

topology based on tributary nodes with the FD&W

PHOTOPTICS 2022 - 10th International Conference on Photonics, Optics and Laser Technology

140

unampliﬁed solution is performed. In this scenario,

the main PLI is the in-band crosstalk that arises due

to the wavelength reuse in these networks. First, we

present the analytical formalism used to assess the

impact of this phenomenon, based on the electrical

SNR at the receiver output. We start our analysis by

presenting the SNR at the receiver output without in-

band crosstalk (Rizzelli et al., 2021), and assess the

maximum number of spans a signal can cross in a

lightpath of the horseshoe topology. After, we intro-

duce the in-band crosstalk in our formalism and as-

sess the respective degradation in terms of the number

of spans.

The SNR without in-band crosstalk is given by

(Rizzelli et al., 2021)

snr =

p

Rx

σ

2

th

P

CW

L

+ P

CW

L

σ

2

nLO

CMRR + σ

2

shot

+

p

Rx

SNR

q

(7)

where P

CW

L

is the continuous wave optical power of

the local oscillator (LO), CMRR is the Common Mode

Rejection Ratio of the balanced photodetector, SNR

q

is an additional term to include several implementa-

tion penalties occurring in a coherent system, σ

2

th

is

the variance of the transimpedance ampliﬁer thermal

noise (TIA), σ

2

nLO

is the variance of the LO transmit-

ter relative intensity noise (RIN) and σ

2

shot

is the vari-

ance of the shot noise. The variances σ

2

th

, σ

2

nLO

and

σ

2

shot

are given, respectively, by (Rizzelli et al., 2021)

σ

2

th

=

i

2

T IA

·B

RX

eq

8·R

2

λ

; σ

2

shot

=

q·B

RX

eq

2·R

λ

; σ

2

nLO

= RIN ·

B

RX

eq

2

, where

R

λ

is the overall responsivity of the coherent receiver

(considering also the passives losses before the pho-

todiodes), i

T IA

is the input-referred noise current den-

sity of a TIA, B

RX

eq

is the the receiver effective noise

bandwidth, q is the electron charge and RIN is the LO

RIN parameter. The parameters used to calculate the

SNR are based on the ﬁtting to experimental results

and are given in (Rizzelli et al., 2021). Only a differ-

ent symbol rate of 26.75 Gbaud is considered.

In Fig. 11, the BER as a function of the received

signal power for different local oscillator powers is

presented, considering QPSK modulation, and ob-

tained from the SNR calculated from Eq. 7. The re-

quired target BER assumed for the system is 4 ×10

−3

(HD-FEC) or 2 × 10

−2

(SD-FEC) (Rizzelli et al.,

2021). As observed in Fig. 11, for each LO pow-

ers of 6, 10, 14 and 18 dBm, the minimum received

power required to achieve the target BER is -36.1, -

35.7, -35.6 and -33.8 dBm, respectively, for HD-FEC

and -38.9, -38, -37.9 and -36 dBm for SD-FEC, which

leads to at least 2.2 dB of power budget improvement

for SD-FEC in relation to HD-FEC.

From Fig. 11, it can also be observed that the

LO power of 14 dBm leads to the minimum received

Figure 11: BER as a function of the received signal power

for different local oscillator powers P

CW

L

, for QPSK signals.

power that ensures the target BER, while P

CW

L

= 6

dBm leads to the higher and most demanding received

power to reach the same target BER. Hence, in the

following results, we consider P

CW

L

= 14 dBm, as

the optimum LO power that leads to the target BER.

So, knowing the minimum received power, the main

goal is to compute the maximum power budget and

the maximum reach of the horseshoe network with an

unampliﬁed solution, considering 10 km spans and a

reference input power of 0 dBm, taken from the maxi-

mum input power considered in (Rizzelli et al., 2021).

The maximum power budget is, therefore, 36.1 dB

and 38.9 dB, respectively for the HD-FEC and SD-

FEC.

Then, the insertion losses per span for hub-

span-tributary, tributary-span-hub and tributary-span-

tributary transmission, as shown in Fig. 12, are com-

puted.

Figure 12: Possible span conﬁgurations considered to eval-

uate the performance of the unampliﬁed solution.

In Tab. 3 are shown, the insertion losses consid-

ering the conﬁgurations of Fig. 12 for different S/C

dimensions are shown. Note that, the green values are

the ones that are feasible for each conﬁguration. As

can be seen in Tab. 3, by comparing the power bud-

gets and the insertion losses of just one span with S/Cs

1:16, it can be concluded that the power budgets are

not sufﬁcient to accommodate these insertion losses.

A possible solution to decrease the insertion losses

per span is to reduce the S/C dimension in the A/D

structures. So, in the tributary nodes shown in Fig.

Impact of Physical Layer Impairments on C+L-band Metro Networks

141

4, the S/Cs 1:16 are replaced by S/Cs with smaller

size, as considered already in some works (Emmerich

et al., 2020b), (Emmerich et al., 2020a), for the un-

ampliﬁed solution. The associated insertion losses are

shown in Tab. 3. The A/D structure of the hub node

remains the same as in the ampliﬁed scenario.

Table 3: Insertion losses considering the conﬁgurations of

Fig. 12 for different S/C dimensions in the A/D structure of

the tributary node.

Total ILs [dB] S/C 1:2 S/C 1:4 S/C 1:8 S/C 1:16

Hub-span-trib. 36.1 39.1 43.1 47.1

Trib.-span-hub 32.1 35.1 39.1 43.1

Trib.-span-trib. 19.7 25.7 33.7 41.7

It can be concluded that one span is possible

with S/Cs 1:2 for the conﬁgurations of Fig. 12,

whereas only the tributary-span-hub and tributary-

span-tributary conﬁgurations are possible for S/Cs

1:4, and, for the S/C 1:8, only the tributary-span-

tributary conﬁguration is possible. Two spans trans-

mission is not possible.

In order to take into account the impact of in-band

crosstalk, the SNR given by Eq. (7) is rewritten as,

snr =

p

Rx

σ

2

th

P

CW

L

+ P

CW

L

σ

2

nLO

CMRR + σ

2

shot

+

p

Rx

SNR

q

+ p

xc

(8)

where p

xc

is the crosstalk power deﬁned as p

xc

= X

c

×

p

rx

, with X

c

the crosstalk level.

In Fig. 13, the BER as a function of the re-

ceived signal power with different crosstalk levels is

shown. The BER without crosstalk is also repre-

sented for comparison purposes. We have consid-

ered P

CW

L

=14 dBm since it leads to the lowest re-

ceived optical signal power. So, for an input signal

power of 0 dBm, and considering the insertion losses

shown in Tab. 3, for S/Cs 1:2, 1:4 and 1:8, when the

crosstalk level ranges between -35 and -25 dB, the

optical power budget is still feasible for all cases as-

suming a BER=2×10

2

(SD-FEC). For X

c

=-20 dB, the

minimum received power is about -37.4 dBm, consid-

ering SD-FEC, leading to 1.5 dB degradation in rela-

tion to the case without crosstalk. Although the opti-

cal power budget is reduced, the unampliﬁed solution

is still feasible for all the conﬁgurations in Fig. 12 for

one span. For a 2 spans transmission, the most fea-

sible conﬁguration is tributary-span-tributary due to

its lower insertion losses, which are very close to the

highest power budget.

Figure 13: BER as a function of the received signal power

with the crosstalk level as a parameter.

5 CONCLUSIONS

In this work, the L-band tributary node architectures

known as FD&W with ampliﬁed and unampliﬁed so-

lutions have been studied analytically. The hardware

components, cost, power consumption, offered capac-

ity and the main PLIs that impair the transmission per-

formance in such nodes have been studied,

When the L-band components cost is twice the C-

band cost, the L-band hub and tributary nodes cost is

3.5 times higher than the corresponding C-band node

cost. The transponders contribute around 99% to the

node ﬁnal cost.

For the ampliﬁed solution, considering an horse-

shoe network with 2 hub nodes, tributary nodes, and

10 km spans, and two worst path cases of signal trans-

mission, the optimum OSNR achieved at the optical

receiver input is above 14 dB, when ASE noise and

NLI are considered. This leaves enough margin to

guarantee the minimum OSNR in the ampliﬁed solu-

tion.

In unampliﬁed solutions, to accomplish the power

budget, smaller S/Cs 1:2 and 1:4 must be used in the

A/D structure of the tributary nodes, and transmission

should not exceed one ﬁber span. This allows fre-

quency reuse in the horseshoe network, leading to in-

band crosstalk. For the crosstalk level of -20 dB, a 1.5

degradation on the optical power budget is observed.

ACKNOWLEDGEMENTS

This work was supported under the project of Instituto

de Telecomunicac¸

˜

oes UIDB/50008/2020.

PHOTOPTICS 2022 - 10th International Conference on Photonics, Optics and Laser Technology

142

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